C650立式機(jī)床電氣系統(tǒng)的設(shè)計(jì)【全套設(shè)計(jì)含CAD圖紙】

資源目錄里展示的全都有，所見即所得。下載后全都有,請(qǐng)放心下載。原稿可自行編輯修改=【QQ：401339828 或11970985 有疑問可加】 A High Performance PWM Current Source Inverter Fed Induction Motor Drive with a Novel Motor Current Control Method Mika Sal0 and Heikki Tuusa Department of Electrical Engineering, Power Electronics Tampere University of Technology P.O.Box 692, FIN-33 101 Tampere, Finland Abstract - This paper presents a high performance vector control- led PWM current source inverter (PWM-CSI) fed induction mo- tor drive where only the measured rotor angular speed and the de-link current are needed for motor control. Novel methods for compensating the capacitive currents of the motor filter and damping the motor current oscillations in the transient conditions are presented. The validity of the proposed methods are verified by simulation. I. INTRODUCTION The rapid development of power and micro electronics in re- cent years allow the use of induction machine also in high per- formance motor drives. At low- and medium power level the variable speed induction motor drives are usually realized us- ing a PWM voltage source inverters (PWM-VSI). However, the switched voltages yield high dddt-voltage slopes over the stator windings, which stresses the insulations and causes bear- ing current ploblems. A possible solution for this ploblem is the use of PWM current source inverter (PWM-CSI) (Fig. 1). Both the voltages and the currents of the machine are nearly sinusoi- dal and therefore the voltage stresses in the machine windings are low. In the PWM current source inverters a C filter has to be in- serted on the load side to reduce the current harmonics. Due to the capacitive currents of the filter the motor current references are not realized accurately, which can be the cause for unsatis- factory performance and instability problems. A few meth- odsl,2, which are based on the measurement of the load capacitor voltages, have been reported to solve the problem. However, with the compined steady state equations of the load filter and motor the capacitive currents can be compensated without any measurements. Line bridge On the other hand, the C filter and the machine inductances form a resonance circuit which is stimulated especially when the motor current references are changed. Some methods3,4, based on the measurement of the motor voltages and/or cur- rents have been proposed to damp the motor current oscilla- tions in the transient conditions. However, in PWM-CSI drives motor current measurements are not needed for protection since the overcurrent can be detected with the dc-link current sensor. So, it is preferred to use control methods where motor current measurements are not needed because in that case the motor current sensors can be totally eliminated. In the present work the control system of the PWM-CSI fed drive is under investigation. The line side converter has been studied earlier5,6 when also the prototypes of 5 kW and 100 kW have been built. The final goal is to develop a high per- formance motor drive with minimum hardware requirements. The proposed vector control system is realized in the rotor flux oriented reference frame. The capacitive currents of the load filter are compensated without any measurements using the combined steady state equations of the load filter and the mo- tor. Also, a new method for damping the motor current oscilla- tions in the transient conditions is presented. The method is based on the combined dynamic equations of the load filter and the motor and does not need any measurements. However, the speed sensor is included to get the drive also to work well near zero speed. 1 1 . VECTOR CONTROL OF THE PWM CURRENT SOURCE INVERTER FED INDUCTION MOTOR DRIVE Fig. 1 shows the main circuit of the PWM current source in- verter fed induction motor drive. Llif and Clif are the induct- ance and capacitance of the line filter and usup the supply Lrlr Load bridee Fig. 1. The main circuit of the PWM current source inverter fed induction motor drive 0-7803-5421-4/99/$10.00 0 1999 IEEE 506 voltage. Clof is the load filter capacitance. The line and load bridges are identical. Both bridges consist of six controllable switches such as IGB transistors (IGBTs). Antiparallel diodes of the IGBTs in the commercial power modules are also shown in the figure. Because of these diodes and very low reverse voltage blocking capability of the IGBTs, additional diodes have to be connected in series with the transistors. A smoothing inductor ( Ldc) is connected between the bridges. In the PWM-CSI drives the line converter is used to control the dc-link current. The function of the line converter is syn- chronized with the supply voltages. By changing the modula- tion index in the line bridge the dc-link voltage, i.e. the dc-link current, can be controlled. In the line-voltage-oriented refer- ence frame the active and reactive power of the line converter can be simply controlled with the real and and imaginary axis components of the supply current vector5,6. The line filter takes reactive power which can be compensated by the control system5,6. The stator currents are generated by the load converter. The load filter takes capacitive currents which are proportional to the square of stator frequency in the constant torque region and linearly proportional to the stator frequency in the field weak- ening region. A. Rotorflux based vector control system In the vector control strategies the AC motors are controlled like dc motors which have independent channels for flux and torque control. Fig.2(a) shows the vector control system which is realized in the rotor-flux-oriented reference frame and is based on indirect vector control scheme7. It should be noted that the control system of the line converter is not shown in the figure. Detailed description of the line converter control can be found in 5,6. The electromagnetic torque of the induction motor in the ro- tor-flux oriented reference frame can be written as 3 L m - t e = 2 -p+$ Lr SY where p is the number of pole pairs, L, magnetizing induct- ance, Lr rotor self inductance, limd rotor magnetizing current and i the imaginary axis component of the stator current vector in the rotor flux based coordinate system. Below nomi- nal rotor speed is kept constant and the electromagnetic torque is controlled with is, of which reference value is the output of the speed controller. Above nominal rotor speed the reference value of the magnetizing current is inversely propor- tional to the stator frequency. lirnr1 can be controlled with the real axis component of the stator current vector is, expressed in the rotor-flux-oriented reference frame, as follows: sY d - T -limd + = is, dt ref dc I Machine Load Converter Soeed Controller + . I+ Trit a) .ref + invy ref SY campy Phase-error compensation eompx .ref + .ref LgX invx b) + invy Referenc filter Oscillations damping Referen filter + .ref ref 8x 1inv.x e) Fi 2 a) Vector control of the PWM-CSI fed drive in the rotor-flux-oriented reference frame. b) Compensation of the motor current phase-error. c) Damp- ing of the motor current oscillations. In the indirect vector control system the rotor flux angle is calculated as a sum of the measured rotor angle and the refer- ence value of the slip angle in the following way: :ref e , , = er+ - sy dt T :I (3) If the angular rotor velocity w, instead of 8, is measured, as is the case in Fig. 2(a), (3) can be written as (4) Rotor flux angle is needed to transform the inverter current ref- erence vector ; ; : $ to the stationary coordinates &$. Su- perscript mr refers to the rotor flux based reference frame. In the proposed vector control system only the rotor angular speed and the dc-link current measurements are needed for mo- tor control. The measured dc-link current is needed for the modulator realization 8,9 in both converters and for dc-link current control in the line converter. The dc-link current refer- ence value is generated in the load converter as follows: 507 where the constant c 2 1 i.e. the magnitude of the dc-link cur- rent should be equal or greater than the length of the inverter current reference vector in order to keep the modulation in the linear region. B. Compensating the motor current phase-error The problem in Fig. 2(a) control system is that the stator cur- rent reference vector is not realized accurately because of the capacitive currents of the load filter. With the combined steady state equations of the load filter and motor the capacitive cur- rents can be compensated without any measurements. Next, the equations needed for compensation control are derived. The stator voltage equation of the induction motor in the sta- tionary reference frame can be expressed as where o is the resultant leakage constant. The load filter ca- pacitor voltage can be written as (7) and the load capacitor current as ilofc = iinv- is. (8) By substituting (8) into (7) and the resulting equation into (6) (Es= Elofc) following expression is obtained: d i d imr ( i n V - , ) d t =RSis+oL -+(I-o)L - (9) Sdt Sdt When (9) is expressed using the quantities of the rotor flux-ori- ented reference frame we have -mr By solving (10) for iinv the following equation is obtained: -mr di, -mr m r z a m r is where w mr = &,/dt. According to (1 1) the effect of load fil- ter in steady state can be compensated as follows: When (12) is expressed in terms of direct and quadrature axis components we have and 2 .ref = R c 0 jref .ref compy s lof mr sx -(3Lsclofwmrsy (14) L where the reference values of the stator current components and the rotor magnetizing current are used. In the constant flux region I imrl = is, and (13) can be written as .ref ref 2 .ref compx = -RsClofWmrisy -Lsclofw mrsx (15) The proposed compensation method is shown in the block diagram form in Fig. 2(b) which replaces the area surrounded by the broken line in Fig. 2(a). C. Damping the motor current oscillations The load filter capacitance and the machine inductances form a resonance circuit which is stimulated especially when the motor current references are changed. One solution to over- come this problem is to use combined dynamic equations of the load filter and the motor. By taking into consideration the dynamic terms of the stator current vector in (1 1) we have -mr 7-mr The dynamic terms of rotor magnetizing current are not in- cluded in (16) because limrl changes much more slowly than .mr L and also because the rotor magnetizing current is normally kept constant. When (16) is expressed in terms of direct and quadrature axis components we have and 508 However, because in practice real stator currents cannot follow step responses of the supply current references, modified (fil- tered) current references( z;lf and isryef) are used in (17) and (18). The proposed damping method is shown in block diagram form in Fig. 2(c), which replaces the area surrounded by the broken line in Fig. 2(a). An example of filtering the stator cur- rent references in discrete case is shown in Fig. 3 where a change in stator current reference value is obtained at time tk. The realization of the reference value is begun at time tk+l be- cause of the one time interval calculation delay. After that the orginal reference value is realized during four time intervals. For microconpoller implementation (17) and (18) have to be discretized when we have At - .k + 1 ldampx= RsClof -ref At Ais, k + 1 + oLsCl0f L At k + l (19) and k+l -ref k + 1 -ref k = ( A * - % )/At (22) In discrete realization the average values of modified stator current references during a time interval should be used in the summing point shown in Fig. 2(c). These can be expressed as :ref,k+1 :ref,k+l -ref,k+2 lsxy,av = (zsxy + isxy )/2 (23) Finally, the modified current reference, which is filtered ac- cording to Fig. 3, can be written as SXY ref,k+l - i -ref,k+2 - :ref,k+l isxy - zsxy + 0.25 ( isxy ref, k . ref, k - 1 ref, k - 1 - irefk-2) (24) ) + 0.3 ( isxy SXY + 0.45 (isxy - zsxy and r i k + 1 1 1 1 . SIMULATION RESULTS L J where (both components combined in one expression) -ref,k+2 :ref,k+l = ( isxy - xy /At % . ref tk tk+l tk+2 tk+3 tk+4 tk+5 Fig. 3. Example of filtering the stator current references. The simulation is based on the parameters shown in Table I. However, due to the skin effect the stator resistance in the res- onance frequency of the load filter (360 Hz) is considerably larger than that shown in the table. Therefore, three times the value given in the table has been used in the simulation model. The model has been built in discrete form to have close analogy with the future microcontroller implementation. The model has been built using per unit values. The base values are: current io= A, voltage uo= ./zUs, angular speed wo= 21150/p, flux wo= U,/ (21150) and torque to= (3/2) uoio/wo. Fig. 4 shows simulation results of the proposed damping method where the y-axis component of the stator current refer- ence vector is suddenly changed. The stator current references are filtered as shown in Fig. 3. The discrete time interval At in (19)-(24) is 200 ps. Fig. 4(a) shows the phase-A stator current when the damping method is not used. Fig 4(b) shows the sim- ulation result when the damping method is used. It can be seen that with the proposed damping method the current oscillations can be considerably reduced. Fig. 5 shows the simulation results of the entire vector con- trol system. The reference values are shown with the broken line and realized values with the solid line. The proposed con- trol methods of compensating the reactive power drawn by the load filter and damping the stator current oscillations are used. The magnetization of the motor is beginned at 10 ms. The ref- erence value of the rotor flux is rate limited in order to keep iLE* at an acceptable level. The final value of the rotor flux in the constant flux region is set to 0.9 p.u. Then, at 100 ms the 509 TABLE I SIMULATION PARAMETERS Nominal Stator phase Voltage us 230 v Nominal stator current 1 , 16 A NOmindl Shaft power PN 1.5 kW Magnetizing inductance L, 80 m H Stator leakage inductance L,I 4.5 mH Rotor leakage inductance L,I 4.5 mH Stator resistance 50 Hz Rs 0.6 Q Rotor resistance 50 Hz R, 0.7 Q Inertia moment J 0.1 kgm2 Friction constant B 0.01 Nms Nominal speed nN 1440 r/min Load filter capacitance CI, Line filter capacitance Clif Line filter resistance50 Hz Rlif Number of po:e pairs p 2 22.5 pF 22.5 pF 1.2 mH 0.1 Q Dc-link inductance Ldc 20 mH Line filter inductance Llif I 0 0.05 0.075 0 025 f , r n e ( s ) I I 0 0.05 0 075 07 ,me(=) 0 025 Fig, 4. Simulated waveforms of phase-A stator current at sudden change in ig&. a) Without oscillation damping control. b) With oscillation damping control. - I / I 0 2 0 3 0 4 I 0 2 0 . 3 0.4 tirne(s) 0.1 Fig. 5. Simulated waveforms of PWM current source inverter fed induction motor drive. a) Rotor flux. b) x-axis component of the stator current. c) Angular rotor speed. d) y-axis component of the stator current. e) Electromagnetic torque. f) dc-link current. g) Phase-A stator current. h) Phase-A stator voltage. 510 reference value of the angular rotor speed is changed from 0 to nominal value (0.96 P.u.) and at 250 ms to 0.7 p.u. Finally, at 350 ms the load torque is changed from 0 to nominal value (0.71 P.u.). The reference value of the dc-link current is caluc- ulated using c=1.2 in (5). Negative slopes of the dc-link current reference value are filtered using time constant of 100 ms. The output of the speed controller is limited to 1.5 p.u. According to FigS(b) and 5(d) it can be seen that the stator currents follow the stator current references well and that the oscillations in the stator current are low. I21 31 41 IV. CONCLUSIONS In this paper, the control of the PWM current source inverter fed induction motor drive has been discussed. The control sys- tem has been realized in the rotor-flux-oriented reference frame. New methods for compensating the reactive power drawn by the load filter and damping the stator current oscilla- tions without any measurements have been presented. The tests with simulation model show excellent performance in both steady and transient state conditions. REFERENCES I J. Cambronne, B. Semail and C. Rombaut, “Vector control of a P. W.M. current source inverter-fed induction nioto?, 4th European Conference I51 161 on Power Electronics and Applications, Firenze, Vol. 2, pp. 177-181, 1991. A. Dakir, R. Barlik, M. Novak and P. Grochal, “Computer simulations for two angular-speed-control systems o f a current source inverter feeding an induction machine”, Proceedings of the IEEE International Symposium on Industrial Electronics, pp. 940-945, Vol. 2, 1996. D.-C. Lee, D.-H. Kim and D.-W. Chung, “Control of PWM current source converter and inverter system for high performance iizduction motor drives”, 22nd International Conference on Industrial Electronics, Control and Instrumentation, Vol. 2, pp. 1100-1105, 1996. P. Eichenberger and M. Junger, “Predictive vector control of the stator voltages for an induction machine drive with current source inverter”, 28th Annual IEEE Power Electronics Specialists conference, pp. 1295- 1301, Vol. 2, 1997. M. Salo and H. Tuusa, “A current-source PWM-rectifier with reactive power control”, 4 t h European Power Quality 97 Conference, Nurn- berg, pp.159-168, 1997. M. Salo and H. Tuusa, “Active and reactive power control o f a current- source PWM-rect$er using space vectors”, FINPIE 91, Espoo, pp.75- 78, 1997. P. Vas, Electrical machines and drives, Oxford, Oxford university press, pp 45-50, 1992. H. 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